System for Preventing Transformer Saturation

ABSTRACT

System and method for managing a cumulative DC offset in a magnetizable material. A primary driving AC voltage and a magnetic flux sensor. The flux sensor output is continuously received into memory while the flux sensor output for each phase half-cycle is processed to continuously compute and re-compute in real time a flux-second integral for each half-cycle. The two half-cycle flux-second integrals are compared to each other for a DC offset value and the offset value drives a slow loop DC compensation circuit to steer a PWM control.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.18/164,541 filed Feb. 3, 2023 which is a Division of U.S. patentapplication Ser. No. 17/101,990 filed Nov. 23, 1920 now issued Mar. 7,2023 as U.S. patent Ser. No. 11/601,041 which is a Continuation-in-Partof U.S. patent application Ser. No. 15/859,656 filed Dec. 31, 2017 nowissued Nov. 24, 1920 as U.S. patent Ser. No. 10/848,086, which claimedpriority to U.S. Provisional Patent application 62/441,329 filed Dec.31, 2016, all of which are hereby incorporated by this reference as iffully set forth herein.

TECHNICAL FIELD

This disclosure relates to controlling magnetic flux density in a volumeby modulation of the electromotive force affecting that volumeproportionate to the measured magnetic flux density within the volume;more particularly it relates to transformer core saturation and methodsof limiting or preventing such saturation; more particularly, it relatesto feedback systems employing direct magnetic flux sensors.

BACKGROUND

AC line voltage regulators maintain a constant output voltage (withinlimits) while the line voltage input to them changes. One popular classof electronic voltage regulators utilize a series injection transformer,the secondary of which is connected in series between the input voltageterminal and the output voltage terminal, and the primary of which isconnected to a switch mode power supply (SMPS) that can supply amodulated in-phase or counter-phase voltage derived from the inputvoltage terminal. A control system monitors the input and outputvoltages and commands the SMPS depth of modulation and phase so that thetransformer secondary voltage boosts or bucks the input voltage in orderto maintain a constant output voltage.

At full boost and at full buck, the primary of the series injectiontransformer is essentially driven directly from the input terminalvoltage and this, the maximum possible high line voltage, becomes acritical factor in the design specification for the series injectiontransformer.

Economics and efficiency both dictate that the transformer be as small,light, and quiet as possible, leading conventionally to the use of atransformer with a toroidal wound steel core rather than with a coremade of stacked steel laminations. This is because, magneticallyspeaking, a toroidal core appears to be largely magnetizable steel inseries with a very small non-magnetizable air gap, while the stackedcore has a significantly smaller ratio of steel to air gap. This allowsa toroidal transformer to be smaller and more efficient than astacked-core transformer for the same power rating. However, thereduction in the ratio of air to steel leads to more abrupt saturationof the core.

The most desirable transformer steel alloys are able to be magnetized toa high maximum flux density, meaning that a transformer core with thesealloys can be smaller than one made of a low maximum flux density steelto reach the same total flux. Also high flux density steels typicallyrequire less energy to be magnetized in both directions reversibly overan AC cycle, meaning that the transformer will be more efficient and runcooler than one with ‘lossy’ steel.

Such high flux density alloys do tend to saturate abruptly at theirdesign flux limit however, and at saturation the magnetic field in thetransformer core no longer increases in direct proportion to the currentin the transformer primary winding. In an unloaded transformer, theprimary current is referred to as the core excitation current. It is therate of increase in the core magnetic field that limits the rate of riseof excitation current and at saturation this magnetic field rate of risereduces to almost nothing. Thus at saturation the excitation currentincreases rapidly and can reach a level that is limited solely by theresistance of the primary winding.

In other words, a saturated transformer draws very high currents similarto a short circuit. In fact this rise in primary current hashistorically been used as a 2nd order, or indirect, indicator of coresaturation, but it is difficult and/or impossible to separate thisindication from load related, over-current events due to thenon-linearity of transformers.

A second critical factor in the design specification for thetransformer, of equal importance to the maximum excitation voltage, isthe frequency of the AC line. This is because the transformer coremagnetic field is driven in opposite directions for the duration of eachhalf-cycle of the AC voltage, and theoretically the magnetic fieldre-centers each cycle, assuming the half-cycle time durations andvoltages are equal. Smallest and most efficient transformers aredesigned to be driven just to the edge of saturation on each half-cycle.Thus the cores in such transformers are very nearly in saturation twiceper full AC cycle. The magnetic flux peak in the core material comesjust at the end of each half-cycle, when the integral of Volts X Timereaches a peak. This Volt-Second integral can be thought of as theaverage voltage across the transformer primary terminals over time. ThisVolt-Second integral is unique to each transformer design, and thetransformer will saturate if this number is exceeded. The magneticcharacteristics of any given transformer core are typically describedgraphically in the form of a B-H curve, as will be appreciated by thoseskilled in the art.

If power to the load on a power line remain constant, the Volt-Secondintegral of the AC supply should be zero over the long term. In simpleterms, this means that the voltage on every positive half-cycle isexactly equal to the voltage on every negative half-cycle for long timeperiods, ie that the AC waveform is voltage symmetrical.

However, line loading is almost never constant and the Volt-Secondintegral can be perturbed for various periods of time by a multitude ofcauses (such as by SCR controlled loads), sometimes to the point ofcausing long term transformer saturation with consequent equipmentoverheating and outages due to circuit breakers or fuses opening, any orall of which can result in taking equipment offline. Also the pulsedcurrents caused by the saturation can disturb sensitive equipment thatshares the same utility line by degrading the line voltage regulation.

In addition, it is not just long-term AC waveform asymmetry that is aproblem; it is also a problem at shorter time scales. The Volt-Secondintegral begins averaging on the instant that a transformer is connectedto a power source. It is believed that transformer steel has a kind of‘flux memory’. If the transformer was disconnected at the instant whenthe core was near saturation in either direction, it would maintain aremnant magnetic field in that same direction. When power is re-applied,depending on the instantaneous AC polarity, the remnant field will addto or subtract from the field being induced by the applied voltage. Ifthe fields add, the core can go into deep saturation long before theVolt-Second integral number is reached and can stay saturated for aconsiderable portion of a half-cycle, meanwhile drawing a very largecurrent. This is why toroidal transformers generally require currentlimitation circuitry to protect equipment.

To summarize: small, light and efficient torroidal transformers have anincreased tendency to saturate. Saturation is detrimental to equipmentlife and good supply voltage regulation. A transformer is most efficientwhen it is operated at high flux levels, ie just below the saturationpoint. Therefore there is value in a system that can reliably control atransformer to run near its peak flux rating by sensing and limiting theflux fast enough to prevent saturation.

To take action to prevent saturation, the flux state of the transformercore must be known. Various schemes for doing so have been proposed,including E-core with air gap in an external leg (Patel 1980), reducedcross-section and additional magnetic path (J. A. Ferreira 1997),integration of applied voltage by external RC network (D. Wilson 1981),measurement of magnetizing current (J. W. Kolar 2000), superimposedorthogonal flux density with external coil (S. Cuk 1982) and sharedmagnetic path between a main core and an auxiliary core, where magneticflux density through the main core changes properties of the sharedmagnetic path, modifying, for example, the inductance seen from theauxiliary core winding (G. Ortiz et al, undated Swiss Power Institutepaper).

It is known that core flux state can be estimated by applying a knownalgorithm to the measured excitation voltage over time to calculate fluxstate from the model inside the algorithm. However, algorithmic meansare only good if the model is detailed, precise and accurate, and suchmeans require a rather long history of the conditions in which thetransformer is operated before the algorithm becomes reliable. In otherwords, the algorithm is nearly useless when the transformer is firstswitched on. Crafting a good model is also a difficult task, a task thatmust be repeated for every different transformer to which the model isto be applied.

Alternatively a means to directly or indirectly measure the flux densityof the core magnetic field can be applied.

Direct measurement of the flux state of the transformer core hasrecently been done either with Hall-effect devices inserted intoapertures in the core, or by measuring the permeability of a portion ofthe core. Indirect measurement of the flux state of the transformer corecan be done using a secondary core structure and then inferring the fluxlevel from the scaled permeability.

Hall-effect devices have a relatively small signal to noise ratio, alimited temperature range, and a tendency to drift. Permeabilitymeasuring circuits and secondary core structure can be complicated, slowto respond relative to the flux change in the main core and oftenrequire individual calibration because of difficulty in preciselyrepeating the mechanical placement of the sensing element.

Additionally, transformer core flux naturally alternates (swings) frompositive to negative following the AC voltage input, and if thevolt-second integral of the drive voltage is not balanced (from a DC, orhalf-cycle, perspective), the core can exhibit a DC offset (meaningthere is higher peak flux density in one half cycle than during theother). This DC offset can exist for long periods of time (increasingthe core losses) without it actually saturating the transformer.

Since half-cycle saturation can occur in very small orders of time, ienear instantaneous, what is needed is a means both to directly andprogrammatically (automatically) detect and correct (limit) magneticflux densities and DC offset irregularities in equally small orders oftime commensurate with the dynamics of such saturation.

DISCLOSURE

A relatively recent technological innovation, the GiantMagneto-Resistive (GMR) (also sometimes referred to as‘magneto-resistant’) sensor is applied in a surprising and novel way toyield direct, 1st order, near instantaneous, programmatic measurementand limitation of the core flux and or DC offset irregularities.

The GMR is small enough to be easily positioned within the transformercore material; for example within in a small bore hole drilled into thecore, and this bore causes only minimal disruption to the flux path.GMRs are small, robust, fast, heat tolerant and have a relatively largesignal to noise ratio for the range of fluxes being measured. GMRs areavailable in a bridge topology, so the drift over a large temperaturerange is not a great problem. Alternative magneto-resistive technologiesmay also be employed, such as Tunnel Magneto-Resistive (TMR) or othermagnetic tunnel junction based sensors including giant magnetoresistance(GMR), colossal magnetoresistance (CMR), and extraordinarymagnetoresistance (EMR). These sensors are employed as otherwisedisclosed herein for GMR.

The disclosed system limits the magnetic flux density in a transformercore below a level at which the core is in saturation by modulation ofthe magneto-motive force produced by the transformer primary winding,the modulation being controlled by direct, real-time (nearinstantaneous) and continuous measurement of the flux density within thecore. ‘Direct measurement’ as used in this disclosure means measurementfrom within the core by a sensor placed within the core.

The disclosed system uses a GMR to directly sense flux density and thenuses the GMR output signal to input and drive various firmwarecomputations which in turn modify the voltage at the primary just beforesaturation occurs. Any line transients resulting from these half-cyclemodulations are desirably removed or otherwise conditioned in ways thatwill be known to those skilled in the art.

Using the sensor output as a threshold measurement for limitingexcitation voltage as herein disclosed is only one possible use of thedisclosed sensor. In an alternative or possibly parallel implementationof the sensor, the sensor output is integrated over time to compute theequivalent of a ‘flux-second’ value. This facilitates optimal control ofthe magnetic flux by continually computing in real time the effect onthe magnetic core flux of the volt-seconds applied to the controlwinding. These computations are then used to programmatically andautomatically regulate control winding voltage and thus keep the fluxconstantly centered (or very nearly so) on the B-H curve.

It will be appreciated that the flux-seconds value does not quite have alinear relationship to the volt-seconds value as applied to the drivewinding. They are not generally aligned, especially with reference tothe time domain. This nonlinearity with respect to time goes beyond thetypical 90 degree lag in the flux density mentioned herein. Considerthat a change in the DC offset of the drive voltage can take someseconds or more to manifest an effect of that offset in the core fluxbalance. Steerage of the core flux is rather “squishy” compared to achange of volt-seconds input on the drive winding, due at least in partto variable latencies, which in turn depend on properties of the corematerials. Thus integrating volt-seconds is not the same as, or asubstitute for, a direct first order measurement of flux-seconds.

One suitable amplifier circuit for use with bridge-configured GMRsensors is illustrated in FIG. 1 . This circuit buffers the relativelyhigh impedance GMR sensor so that it can be remotely located from thebalance of the signal processing circuitry. It is desirable to separatethe flux sensor itself from the balance of the signal processingcircuitry in electrically noisy environments, or for isolationrequirements, or in other physical circumstances, such as will occur tothose skilled in the art.

FIGS. 3 & 4 show the FIG. 1 circuitry as a block labeled “Flux Sensor”.At a minimum the GMR flux sensor, and advantageously the entirecircuitry shown in FIG. 1 , are desirably located within the laminationstructure of the transformer core. This can be accomplished by drillinga small hole, advantageously perpendicular to the laminations, orotherwise by creating a void, either between or within the laminations,and large enough to accommodate the flux sensor. FIG. 7 illustrates animplementation of these circuits for a series injection topology.

A microprocessor and set of instructions for use with the GMR sensoroutput is also disclosed. Use of a microprocessor allows for a low-cost,low-component-count method to condition the sensor output per therequirements of the regulator pulse width modulation (PWM) of the SMPS.Particular instructions for the microprocessor will vary according todesired or required signal levels, timing requirements, power supplyavailability, and the like of a particular physical and electricalapplication, as will be appreciated by those skilled in the art. But apart of all such instruction sets for the disclosed flux sensor includethe following steps and continuous real-time flux saturation monitoring,as illustrated in FIG. 8 . Reading core flux levels continuously and inreal time via the GMR output and then modifying the PWM (such as, butnot limited to, gating it off), for flux levels above a selectable levelvalue during a particular driving voltage half-cycle, are also part ofthe microprocessor instructions.

In FIG. 7 a microprocessor is advantageously employed to provide signalconditioning from the GMR flux sensor and to gate the regulator PWM as afunction of continuous and real-time direct-sensed flux level orsaturation from the flux sensor (thereby limiting the Volt-Secondintegral). The microprocessor also desirably runs, on a half-cycle byhalf-cycle basis, a test of the integrity of the GMR sensor circuitryand provides a hold-off delay, as further discussed herein. The hold-offdelay is selectable and proportional (generally inversely), as will beappreciated by those skilled in the art, to the system frequency.Testing the GMR sensor circuitry includes the step of back-driving thesensor signal lines in various combinations to determine if one of atleast three sensor failure conditions exists: (1) the sensor leads areshorted together; (2) the sensor leads are open; or (3) the sensor leadsare shorted to either power supply rail.

The Hold-Off delay may also be effected by means other than a circuittesting, such as by a programmable 1-2 millisecond delay (desirablyabout 10% of the length of the voltage half-cycle) on the PWM gating atthe beginning of each half-cycle.

Transformer excitation polarity is an input to the microprocessor and achange to polarity triggers the sensor test period (or otherprogrammable hold-off delay) at the beginning of each half-cycle, a timeperiod during which saturation rarely occurs. If there is a sensor faultit is indicated to the supervisor control circuitry by one or more ofvarious means, for instance, a pair of short pulses coordinated withevery drive voltage polarity change. Since a sensor fault disablesnormal flux level monitoring, it requires timely intervention, and mayresult in a predetermined modification of the SMPS PWM.

During each half-cycle flux sensor testing the PWM gating function isturned off, for example, for approximately 10% of the half-cycleduration (about 1-2 milliseconds in a 50 or 60 cycle system) after thetransformer excitation voltage changes polarity. This Hold-Off intervalperiod, during which the flux level is likely not yet affected by thereverse in primary voltage polarity, is believed to contribute to thetransformer efficiently recovering from near saturation, and isdesirable to prevent false triggering of saturation signals after thesepolarity change points. A false saturation signal at the beginning of ahalf-cycle would interrupt the PWM voltage for the remainder of thehalf-cycle, and that would not allow the transformer flux to re-center.

If the drive voltage is conceived, by way of illustration, in anear-perfect transformer to be a square wave of equal positive andnegative durations and that the amplitude is microscopically less thanthat required to saturate the transformer, then the flux level is atriangular wave with flux peaking exactly at the moment when the drivevoltage changes sign, crossing through zero at the midpoint of thehalf-cycle, and reaching an opposite polarity peak at the next voltagesign change.

If the drive voltage then increases just enough so that the flux levelreaches saturation threshold one millisecond before the end of eachhalf-cycle, then the microprocessor interrupts the primary voltage(gates the PWM off) and holds it off for the remainder of thehalf-cycle. The transformer drive decreases or even drops the voltage tozero for the remainder of the half-cycle and the transformer fluxessentially freezes at the threshold level during that half-cycle. Atthe next drive voltage zero crossing the PWM again becomes active, andthe transformer flux begins to be driven toward its opposite polarityand toward the opposite threshold. After sufficient time, in thisexample flux levels balance out and the flux waveform is a triangle wavewith opposite peaks flattened.

Without a Hold-Off Interval (and or if there is noise in the flux sensorsignal in the moment just after the drive voltage changes from positiveto negative) the saturation threshold can be tripped, the PWM made zero,and the transformer drive voltage goes to zero. The transformer fluxtherefore stays almost exactly at the positive threshold for the timeduration of the missing negative half-cycle, then resumes increasing atthe moment the positive drive voltage is re-applied. In this event analmost instantaneous deep saturation occurs.

With the Hold-Off Interval however, the primary winding drive voltagealways has sufficient time to force the flux below the threshold levelso that the desirable PWM gating generally only occurs near the end of ahalf-cycle. Thus ON-time for each drive voltage half-cycle is servo-edto balance out the flux amplitudes in the transformer and thereby tokeep OFF-times to a minimum.

Controlling the transformer saturation by forcing the drive voltage tozero to prevent transformer saturation does also introduce a generallysmall but measurable transient into the transformer output voltagewaveform at the AC zero crossing point. Also it is a reactive ratherthan a proactive means to handle long-term DC offset (AC asymmetry).

An addition to the microprocessor instruction set provides for thelinear GMR sensor output to cause adjustment of the voltage source PWMon a half-cycle-by-half-cycle basis so that any transformer drivevoltage asymmetry is cancelled without disruption. See FIGS. 3, 5 and 7. This technique actively centers the transformer flux excursions andresults in reduction of transformer heating and line current pulsation.Asymmetry of the series injection transformer primary drive voltage istypically the result of differences in the resistance of individualcomponents in the SMPS switching circuit and cumulative PWM timingerrors. Asymmetry of the AC line can also be transmitted to the SMPS andaffect the drive voltage.

A process that equalizes transformer flux excursions over timedifferentially modulates the amplitude of positive vs negativehalf-cycle drive voltages to the series injection transformer. This isdone by multiplying the PWM for positive half-cycles with a gainvariable Kpos which is different from the negative half-cycle gainvariable Kneg. The ratio of these two variables (Kpos to Kneg) can beadjusted by slowly changing the ratio in one direction until, forexample, the transformer begins to saturate on the positive half-cycles,then changing the ratio in the other direction until the transformerbegins to saturate on negative half-cycles. The ideal setting for theratio will then be centered between the values that caused theopposite-polarity saturations. This ratio-setting desirably runscontinuously as a background real-time process or alternativelyperiodically as determined by power line conditions.

It is to be noted that application of the disclosed system is notlimited to flux sensing, flux limiting and balancing applications forseries injection transformers. All transformers and inductors behavesimilarly with respect to flux in their magnet core.

These methods of modulating the PWM and thereby modulating thetransformer primary voltage are advantageous over earlier attempts tocontrol transformer flux excursions by using a separate (additional)transformer winding to inject DC current. However, in the absence of PWMmodulation of the transformer primary, the disclosed system can still beused to control a separate (additional) transformer winding to inject DCcurrent, as is otherwise well known. See circuit example illustrated inFIG. 4 . This is a diagram of a transformer flux centering controlsystem that uses current feedback via an auxiliary transformer winding.This control system senses the transformer core flux levels over timeand uses that information to control current in an auxiliary transformerwinding in such a manner as to maintain the averaged flux level at zero.The auxiliary winding is there not to sense flux, but rather to injectDC current in order to alter the flux level of the core. This actioncompensates for DC offset on the AC line to the transformer.

An electrically isolated, self-powered flux sensor is also disclosed,for example, as an optically coupled flux sensor. In the case ofapplication in higher voltage distribution and power transformers, theflux sensing circuit is altered such that it is a self-powered, andelectrically isolated, flux sensing module. This module triggers anappropriate flux limiting method using some electrically isolatedsignaling method such as, but not limited to, fiber optics. An exampleis schematically illustrated in FIG. 6 . It is desirably housed in anappropriate hermetically sealed container for use in oil filledtransformers.

In some cases, the flux sensor output is used not only as a threshold,but also (or instead) as an input to a processor with programmedinstructions to integrate this output over time. This instantaneous,continuous integration yields what may be called a flux-second valuewhich is analogous to the commonly understood volt-second integration.Since the flux sensor signal is uni-polar, the flux-second value iscoordinated with an AC voltage polarity input in order to assign theflux-second value a bi-polar value (+ or −) that is synchronized withthe drive winding circuit. It will be appreciated that the AC waveformon the transformer drive winding theoretically precedes the flux densitysignal by 90° in cyclical terms.

Using conventional volt-second integration as a control input to thedrive winding can effect a balance of the positive and negative voltagecycles, which in turn can effect a constant transformer flux balance.But by itself it does not provide any direct information about theexistence or magnitude of a DC bias developing in the core. Volt-Secondintegration has long been used as a 2nd order approximation of thetransformer flux density. Now that 1st order measurement of the fluxdensity is achieved, it also makes sense to integrate it over time andto use that information to “steer” the flux balance and keep thetransformer flux “centered” on the B-H curve (See FIG. 14 ).

Using a sensor-inputted and constantly processor-computed flux-secondintegration value as a control input to control the half cycletransformer drive winding voltage accomplishes three things:

-   -   1. sensing and measuring a changing DC offset condition in real        time;    -   2. correcting the changing DC offset by varying the volt-seconds        applied to the drive winding every half cycle; and    -   3. sensing and controlling DC offset enables designing a        transformer which runs at higher flux levels, which in turn        allows for use of a physically smaller iron core, lower core        losses (for instance, iron losses), and greater dynamic range of        the series injection transformer.

This disclosure generally covering flux sensor placement in a toroidaltransformer core is by way of illustration and not limitation. The fluxsensor system works well in any style of transformer core construction.Furthermore, any medium to high permeability magnetic material willserve, including but not limited to powdered iron cores such as may befound in motors, solenoids, inductors, transformers and the like.

A system for preventing magnetic saturation in a magnetizable materialsuch as transformer cores, inductors and the like is disclosed. Ingeneral, a magnetic flux sensor is disposed within the magnetizablematerial, for example in a bore drilled or let into the material of atoroidal transformer core. The transformer or the like has applied aprimary driving voltage having opposite phase half-cycles. The sensortransmits a sensor output that is continuously received into machinereadable memory and used by a hardware processor such as amicroprocessor that is programed to process, in accordance with a set ofinstructions stored in the machine readable memory, the sensor outputand to also continuously compare in real time the sensor output with astored selectable maximum flux sensor output value. This maximum fluxsensor output value may be empirically determined as will be familiar tothose skilled in the art, or provided by the manufacturer, or in someother way determined, now known or later developed. Responsive to thecomparison of real-time sensor output value to the stored maximum value,the microprocessor either allows, during each driving voltagehalf-cycle, the driving voltage to continue unabated while the sensoroutput remains below the selectable maximum value, or triggers a gate tomodify the driving voltage for the remainder of the half-cycle when theselectable maximum value is reached. This modification can be aprogrammatic reduction in driving voltage or change in polarity, or acomplete cut-off to zero.

The disclosed system may also include an electrically isolated,self-powered flux sensor, with or without fiber optic electricallyisolated signal transmission from the flux sensor to the microprocessor.

A particular feature of the disclosed system is a programmable hold-offtime delay having a programmable duration that is programmably andselectively imposed during one or more of the driving voltagehalf-cycles. The hold-off time delay is advantageously part ofinstructions for a system circuitry integrity check.

A method of preventing magnetic flux saturation in a magnetizablematerial is also disclosed. A magnetic flux is induced in a core by aprimary driving voltage having opposite phase half-cycles and a magneticflux sensor within the core continuously senses and transmits to aprogrammable microprocessor a magnetic flux density value from withinthe magnetizable material. The microprocessor continuously receives thetransmitted magnetic flux density value and compares in real time duringeach driving voltage half-cycle each transmitted magnetic flux densityvalue with a selectable and programmatically stored maximum flux densityvalue. The microprocessor triggers a gate to modify the driving voltagefor the remainder of the half-cycle when the selectable maximum fluxdensity value is reached. This driving voltage modification may be inany of the ways discussed above, as well as in ways which will occur topersons skilled in the art and having an appreciation of the disclosedsystem and methods.

The disclosed method likewise desirably includes a hold-off time delayhaving a programmable duration that is programmably and selectivelyimposed during one or more of the driving voltage half-cycles. Themethod advantageously also includes multiplying a PWM for positivehalf-cycles with a gain variable Kpos and for negative half-cycles witha gain variable Kneg, where Kpos and Kneg have different values.

A method of managing cumulative DC offset (with its attendant corelosses) in a magnetizable material is also disclosed. Like the methoddisclosed above for managing and preventing core saturation, a magneticflux is induced in a core by a primary driving voltage, and the magneticflux sensor within the core continuously senses and transmits to aprogrammable hardware processor a magnetic flux density value fromwithin the magnetizable material. In this alternate method, the hardwareprocessor continuously receives the transmitted magnetic flux densityvalue and integrates it over time to compute a ‘flux-second’ value inreal time during each driving voltage half-cycle. The processor controlsthe half-cycle PWM programmatically to ‘center’ the core flux on the B-Hcurve, such as by employing the foregoing Kpos and Kneg values.

A system for preventing transformer saturation is also disclosed, wherethe system uses a GMR or the like to continuously measure and transmit aflux density value for continuous use as a microprocessor input tocontrol a modification of transformer primary voltage (such as discussedabove) when the transmitted flux density value matches a preselectedflux density value approximating a transformer saturation value. Thedisclosed system for controlling a voltage having opposite phasehalf-cycles also includes a set of microprocessor stored instructionsfor controlling a hold-off delay having a programmable duration that isprogrammably and selectively imposed during one or more of the voltagehalf-cycles.

An alternate system for managing and centering a cumulative DC offset ina magnetizable material is disclosed. The system has a primary drivingvoltage with opposite phase half-cycles and a magnetic flux sensoroperably disposed within the magnetizable material. A hardware processoris operatively associated with a machine readable memory and a set ofinstructions is stored in the machine readable memory. The flux sensorhas a sensor output that is continuously received into the machinereadable memory, and the hardware processor is programed, in accordancewith the set of instructions stored in the machine readable memory, toprocess the flux sensor output for each phase half-cycle to continuouslycompute and re-compute in real time a flux-second integral for eachhalf-cycle.

The two half-cycle flux-second integrals are programmatically andcontinuously compared to each other for an instantaneous DC offsetvalue, and the offset value drives a slow loop DC compensation circuit(of a type and configuration that will occur to those skilled in theart) to steer a PWM control to vary the primary driving voltage in sucha way as to add a DC voltage to the phase half-cycle that is deficientand or to subtract a DC voltage from the phase half-cycle that iscontributing to the DC offset, in effect to continuously andautomatically control the DC offset of the AC primary drive voltage toeffect minimal DC offset.

Optional gain variables Kpos and Kneg are advantageously stored in themachine readable memory and the slow loop DC compensation circuitsteering the PWM for positive half-cycles with gain variable Kpos andfor negative half-cycles with gain variable Kneg to center the DC offsetfor each half-cycle.

An alternate method for managing and centering a cumulative DC offset ina magnetizable material with a magnetic flux induced by a primarydriving voltage having opposite phase half-cycles, and a magnetic fluxsensor disposed within the magnetizable material is disclosed. The fluxsensor continuously transmits to a programmable hardware processor amagnetic flux density value for each half-cycle from within themagnetizable material. The processor has access to a machine readablememory in which is stored a set of programmable instructions. A hardwareprocessor continuously and in real time receives and stores into themachine readable memory the transmitted magnetic flux density value foreach half-cycle, and the hardware processor continuously and in realtime during each half-cycle computes and re-computes with eachsuccessively received magnetic flux density value a flux-second integralfor each half-cycle.

The hardware processor programmatically and continuously compares eachof the two half-cycle flux-second integrals to each other for aninstantaneous DC offset value and using the DC offset value, drives aslow loop DC compensation circuit to steer a PWM control to vary theprimary driving voltage in such a way as to add a DC voltage to thephase half-cycle that is deficient and or to subtract a DC voltage fromthe phase half-cycle that is contributing to the DC offset, in effectcontinuously and automatically centering the DC voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit schematic of an aspect of the disclosure.

FIG. 2 is an oscilloscope screen shot of an aspect of the disclosure.

FIG. 3 is a circuit schematic of an aspect of the disclosure.

FIG. 4 is a circuit schematic of an aspect of the disclosure.

FIG. 5 is a circuit schematic of an aspect of the disclosure.

FIG. 6 is a circuit schematic of an aspect of the disclosure.

FIG. 7 is a circuit schematic of an aspect of the disclosure.

FIGS. 8A and 8B are schematic views of aspects of the disclosure.

FIG. 9 is a flow chart of program instructions of the disclosure.

FIG. 10 is a schematic representation of a hardware processor.

FIG. 11 is a flow chart of program instructions of the disclosure.

FIG. 12 is a flow chart of program instructions of the disclosure.

FIG. 13 is a logic flow chart for an aspect of the disclosure.

FIGS. 14A and 14B are schematic views of aspects of the disclosure.

DETAILED DESCRIPTION

A circuit for use with bridge-configured GMR sensors is illustrated inFIG. 1 . GMR sensing bridge 102 has signal output 101 to themicroprocessor.

The flux sensor also enables the flux in power transformers to staycentered in the presence of any DC offset originating in the AC line.Such a circuit is further illustrated in FIGS. 3 and 5 .

FIG. 3 is a diagram of a transformer flux centering control system thatuses modulation of the AC voltage applied to the transformer primary.The Switch ON:OFF time is modulated as a function of supply linepolarity and transformer flux level on a half-cycle-by-half-cycle timebasis, employing conventional line voltage polarity detector 301,amplifier 302, half-cycle compensation 303, PWM 304 and novel fluxsensor 100, as discussed above.

FIG. 5 provides the same function utilizing a series injection topology.The control voltage is increased or reduced each half-cycle so that thepower transformer Volt-Second interval is altered so as to drive theflux excursions to a minimum, and employs conventional power electronicsblock 405, signal conditioner block 406, series injection transformer408 and novel GMR flux sensor 407.

In both illustrations, these control systems sense continuously and inreal-time the instantaneous transformer-core flux levels and uses thatinformation to instantly and in real-time control the transformerprimary voltage in such a manner as to maintain the averaged flux levelat zero. This action compensates for DC offset on the AC line to thetransformer.

In the case of a series injection voltage regulator, the voltageamplitude and polarity are varied as needed on the primary winding of aseries-injection transformer to achieve the desired output voltagecorrection on the secondary winding. This can be accomplished by usingan inverter or an AC to AC converter with an H bridge.

In FIG. 7 transformer saturation is prevented by modifying the voltagesource driving the series injection transformer primary to either reducethe voltage or cut it off to zero (see voltage modification discussionherein) when the GMR sensor indicates a transformer flux levelapproaching saturation. The reduced or zero voltage is maintained untilthe polarity of the line voltage changes, whereupon the full drivevoltage is re-enabled, advantageously at the zero crossing, and thereversed current in the transformer primary begins driving the flux awayfrom saturation. FIG. 7 illustrates SMPS 702, microprocessor block 701,series injection transformer 703 and novel flux sensor 100.

FIG. 7 also illustrates a transformer flux centering control system thatmodulates the AC voltage applied to the transformer primary. Thistechnique is also illustrated in the scope shots below.

In the scope traces shown in FIG. 2 , Cyan is the Sat line, Magenta isthe AC polarity circuit output, Green is the AC voltage on thetransformer primary and Yellow is the current on the transformerprimary. This is a scope shot of a negative saturation where the cyanline goes low, which interrupts the PWM and consequently the primarywinding drive voltage to the series injection transformer.

Also disclosed is a method of providing a direct, first-order fluxmeasurement of a transformer core or other magnetic structure, in orderto control an independent third winding provided to inject DC currentwith the proper polarity into the transformer core such that it reducesthe instantaneous core flux level or, over a longer time frame, it isused to center the core flux to compensate for DC offset, as shown inFIG. 4 . Novel flux sensor 401 and microprocessor block 402 areillustrated.

In FIG. 6 an alternate schematic of an electrically isolated, fiberoptic flux sensor is shown with flux-powered VCC 601, microprocessorblock 602, fiber optic interface 603, and GMR bridge 100.

FIGS. 8A and 8B are schematic views of the flux sensor installation 10and flux sensor 13 and associated electronics positioning within atoroidal core bore 13 with trailing sensor leads 12. Alternate position11 is also shown.

FIG. 9 is a flow chart of microprocessor program instructions for thedisclosed system. Flux sensor installation 30 has SMPS 33, seriesinjection transformer 32 and flux sensor 31. Flux sensor 31 sends fluxdensity value output to microprocessor block 20, received by signalconditioning 21, then passed to threshold comparison 22, 23. If thereceived flux density value is below the threshold the next value iscompared and so on through the half-cycle. If the sensor-transmitteddensity value is above the threshold, PWM gating/modification 27 istriggered, and when zero cross 26 is detected, hold-off timer 25 istriggered, with optional sensor test 24.

FIG. 10 is a schematic representation of a conventional hardwareprocessor 500 with bus 502, processor 504, memory 506, optionalalternate storage 508, optional interfaces 510, and input/output 512.

FIG. 11 is an alternate flow chart of hardware processor programinstructions for the disclosed system. Flux sensor installation 30 hasSMPS 33, flux sensor 31, and series injection transformer 32 which sendsan appropriate value to AC line polarity and zero cross detection 26(which in turn has respective inputs to PWM control 27, hold off timer25 and half-cycle flux-second integration 41). Flux sensor 31 sends fluxdensity value output to hardware processor block 20, received byoptional sensor test 24 and passed to signal conditioning 21 at whichblock the sensor data is passed to half-cycle flux-second integration 41and or (in systems combining both threshold limitation and DC offsetcontrol) to threshold comparison 22.

From integration 41 a value is passed to slow loop DC compensation 42(and optionally to buffer comparator 43, see FIG. 12 ) and then to PWMcontrol 27.

From threshold comparison 22 a value is sent to above/below thresholdcheck 23 and then to optional hold off timer 25 in coordination withsensor test 24 before it goes to PWM control 27 which in turn controlsthe operation of SMPS 33.

FIG. 12 is an alternate flow chart of hardware processor programinstructions for the disclosed system. Flux sensor installation 30 hasSMPS 33, flux sensor 31, and series injection transformer 32 which sendsan appropriate value to AC line polarity and zero cross detection 26(which in turn has respective inputs to PWM control 27 and half-cycleflux-second integration 41). Flux sensor 31 sends flux density valueoutput to hardware processor block 20, received by optional sensor test24 and passed to signal conditioning 21 at which block the sensor datais passed to half-cycle flux-second integration 41. From integration 41a value is passed to buffer comparator 43 and thence to slow loop DCcompensation 42, which then operates on PWM control 27 which in turncontrols the operation of SMPS 33.

FIG. 13 schematically illustrates an example of a logic flow for thefirmware for the flux-second integration and comparison. The filteredoutput from the flux sensor is desirably sampled as frequently as stateof the art hardware will allow. The sampled flux values are binned intosequential positive & negative half cycle bins using the zero crossingand polarity information derived elsewhere from the line voltage. At theend of each half cycle period, which can be either a voltage half-cycleor a flux half-cycle, the values are integrated and subsequentlybuffered for use by a DC slow loop compensation circuitry so that oversome time window that is advantageously adjustable (probably <1 second),the half cycle PWM is gently adjusted differentially with a selectablegain in a manner such that the net flux-second values net to zero. It isbelieved that details of slow loop gain and decay rate will be familiarto those skilled in the art of control loop theory.

In FIG. 13 illustrates a processing logic 800 for continuous, real timeautomatic flux-second integration and comparison. At Initialization 801all variables are set to zero. Loop 802 provides a wait for zerocrossing before summation begins. At the next zero crossing afterinitialization, the summations start. At first approach to IF juncture803, summation for sample bin 1 commences, with SUM 1 starting at zero.All incoming flux values from the flux sensor are summed at integrationloop 804 from NR=0 to N (a hardware appropriate selectable value storedin memory) until at checkpoint 806 NR=N. The bin 1 summed flux-secondvalue is sent to comparator 810 and at operation 808 processing returnsto start for sampling into sample bin 2. Comparison at 810 begins onlywhen both bins have a value.

At IF juncture 803, SUM 2 starts at zero, and all incoming flux valuesfrom the flux sensor are summed at integration loop 805 from NR=0 to Nuntil at checkpoint 807 NR=N. The bin 2 summed flux-second value is sentto comparator 810 and at operation 809 processing returns to start forsampling into sample bin 1.

At operation 810, Sum 1 is compared with Sum 2 and the signed differenceis the difference between successive half-cycle flux-second integralsand that signed value is then sent to slow loop DC compensation circuit42 (see e.g. FIG. 12 ) for further processing.

FIG. 14A represents a typical hysteresis loop (B-H curve) of a coreoperating in a balanced condition. During normal operation the fluxdensity values are contained within the lines of the two S-shapedcurves. It is known that if the magnetizing force becomes too high, theflux density increases until it levels out and core saturation occurs.

FIG. 14B illustrates the hysteresis loop of a core that is operatingwith a (for instance) negative DC offset. The B-H curves are shifted tothe right, and the amount of magnetizing force required to saturate thecore in the negative is less than was required in FIG. 14A. Under theseconditions it is known that the positive flux value will never get ashigh as the negative, and therefore positive saturation is not possiblein this condition. When the positive magnetizing force becomes greaterthan the negative magnetizing force, the B-H curve will then move to theleft until it is re-centered and even potentially continue moving to theleft until the core is in a positive (opposite) DC offset condition (notillustrated).

With regard to systems and components above referred to, but nototherwise specified or described in detail herein, the workings andspecifications of such systems and components and the manner in whichthey may be made or assembled or used, both cooperatively with eachother and with the other elements of the invention described herein toeffect the purposes herein disclosed, are all believed to be within theknowledge of those skilled in the art. No concerted attempt to repeathere what is generally known to the artisan has therefore been made.

In compliance with the statute, the invention has been described inlanguage more or less specific as to structural features. It is to beunderstood, however, that the invention is not limited to the specificfeatures shown, since the means and construction shown comprisepreferred forms of putting the invention into effect. The invention is,therefore, claimed in any of its forms or modifications within thelegitimate and valid scope of the appended claims, appropriatelyinterpreted in accordance with the doctrine of equivalents.

We claim:
 1. A method for preventing magnetic flux saturation in amagnetizable material comprising therewithin a magnetic flux sensortransmitting magnetic flux density values, where the magnetic flux isinduced by a primary driving voltage having opposite and alternatingcurrent phase half-cycles, the method comprising the steps of: amicroprocessor continuously receiving the transmitted magnetic fluxdensity values; the microprocessor comparing in real time during eachdriving voltage half-cycle each transmitted magnetic flux density valuewith a selectable and programmatically stored maximum flux densityvalue, and the microprocessor triggering a reduction of the drivingvoltage to a voltage value greater than zero for the remainder of thehalf-cycle when the selectable maximum flux density value is reached inthe half cycle.
 2. The method of claim 1 wherein the magnetizablematerial is a transformer core.
 3. The method of claim 2 wherein themagnetic flux sensor is disposed within a bore in the transformer core.4. The method of claim 1 wherein the magnetic flux sensor is selectedfrom the group of magneto-resistive sensors including a GMR, a TMR, aCMR and an EMR.
 5. The method of claim 1 further comprising anelectrically isolated, self-powered flux sensor.
 6. The method of claim5 further comprising fiber optic signal transmission from the fluxsensor to the microprocessor.
 7. A system for managing a cumulative DCoffset in a magnetizable material comprising therewithin a magnetic fluxsensor transmitting magnetic flux density values, where the magneticflux is induced by a primary driving voltage having opposite andalternating current phase half-cycles, the system comprising: amicroprocessor operatively associated with a machine readable memory anda set of instructions stored in the machine readable memory; the machinereadable memory continuously receiving the transmitted magnetic fluxdensity values; the microprocessor programed to process, in accordancewith the set of instructions, the flux sensor output for each phasehalf-cycle to continuously compute and re-compute in real time anupdated flux-second integral for each half-cycle; the microprocessorprogramed to compute, from the updated flux-second integral for eachhalf-cycle, an instantaneous DC offset value; and the DC offset valuedriving a slow loop DC compensation circuit to steer a PWM control tovary the primary driving voltage in such a way as to add a DC voltage tothe phase half-cycle that is deficient and or to subtract a DC voltagefrom the phase half-cycle that is contributing to the DC offset.
 8. Thesystem of claim 7 wherein the magnetizable material is a transformercore.
 9. The system of claim 8 wherein the magnetic flux sensor isdisposed within a bore in the transformer core.
 10. The system of claim7 further comprising an electrically isolated, self-powered flux sensor.11. The system of claim 10 further comprising fiber optic signaltransmission from the flux sensor to the microprocessor.